Electronic circuitry

ABSTRACT

Electronic circuitry comprising operational circuits of active switching type requiring timing signals, and conductive means for distributing said timing signals to the operational circuits, wherein the timing signal distribution means includes a signal path that has different phases of a drive signal are supplied via active means at different positions about the signal path where that path exhibits endless electro-magnetic continuity without signal phase inversion or has interconnections with another signal path having different substantially unidirectional signal flow where there is no endless electromagnetic continuity between those signal paths and generally has non-linear associated circuit means where the signal path is of a transmission line nature.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.10/167,200 filed on Jun. 11, 2002, titled “ELECTRONIC TIMING SIGNALCIRCUITRY” which application is itself a continuation-in-part of U.S.application Ser. No. 09/529,076, filed on Apr. 6, 2000, titled“ELECTRONIC CIRCUITRY”, now U.S. Pat. No. 6,556,089, which applicationis a national stage application of international applicationPCT/GB00/00175, filed on Jan. 24, 2000, titled “ELECTRONIC CIRCUITRY,”which international application claims priority to three Great Britainapplications, GB9902001.8, filed on Jan. 30, 1999, GB9901618.0, filed onJan. 25, 1999, and GB9901359.1, filed on Jan. 22, 1999.

The following applications are incorporated by reference into thepresent application:

-   -   U.S. application Ser. No. 10/167,200 filed on Jun. 11, 2002,        titled “ELECTRONIC TIMING SIGNAL CIRCUITRY”;    -   U.S. application Ser. No. 09/529,076, filed on Apr. 6, 2000,        titled “ELECTRONIC CIRCUITRY”, now U.S. Pat. No. 6,556,089;    -   International application PCT/GB00/00175, filed on Jan. 24,        2000, titled “ELECTRONIC CIRCUITRY”;    -   GB9902001.8, filed on Jan. 30, 1999;    -   GB9901618.0, filed on Jan. 25, 1999; and    -   GB9901359.1, filed on Jan. 22, 1999.

This application develops inventive features inherent in U.S.application Ser. No. 09/529,076. One such inventive feature concernscoupling synchronously between traveling electromagnetic waves. Inapplication Ser. No. 09/529,076, endless signal paths exhibitelectromagnetic continuity and afford signal inversion duringrecirculation of those paths, specifically coupling between such signalpaths at positions that sustain and reinforce directionality of signalrecirculations and correlation of their phasing. Suitable signal pathsof a transmission-line nature comprise conductors in parallel relation,usually conductive traces of prescribed dimensions and spacing.

FIELD OF INVENTION

This invention relates to generating electrical signal wave-formsapplicable, but not necessarily only applicable, to use for timingpurposes, including (but not limited to) providing so-called clocksignals on semiconductor integrated circuits.

BACKGROUND TO INVENTION SUMMARY OF INVENTION

As now developed herein, operationally effective directionality andphase correlation is achieved and reinforced for travelingelectromagnetic waves not of wholly recirculatory nature, thus foreffective signal paths that are not electromagnetically endless.

Typically, in implementing this inventive feature, there are multipleinterconnections of signal paths each of a substantially unidirectionalsignal transmission nature, the inter-connections being made plurallybetween nominally phase-correlated positions for signal paths havingdifferent unidirectional signal flows.

In one embodiment, a signal path of a transmission-line nature andhaving substantially unidirectional signal energy flow is coupled toanother signal path which is of a similar nature having oppositesubstantially unidirectional signal energy flow herein referred to as“contra-flow”. Suitable coupling is by active interconnection(s) atposition(s) for phase-correlation of respective opposed signal energyflows. Generally, plural such interconnection positions will have aspaced relation along the respective signal paths, with their spacingscorrelated as periodic intervals relative to signal traverse of thosepaths.

Suitable lay-outs of such transmission-line signal paths can havelocalized adjacencies at their interconnection positions, and can beotherwise spaced to afford areas circumscribed by parts of thecontra-flow signal paths that afford an inversion effect theneffectively similar to that of the endless electromagneticallycontinuous path specifically disclosed in application Ser. No.09/529,076. Suitable such localized adjacencies can be achieved wherethe signal paths or part-paths are of a stepped nature that bringsinterconnection positions close together, typically at tops and bottomsof steps of the stepped paths or path-parts concerned. Ends of pairedconductor type transmission line parts can be loop-connected together toform opposite directions of signal flow paths or path-parts, and/or haveterminations, say at ends of such loop-connected paths or path-parts.

Suitable interconnections can be by way of cross-couplings usingnon-linear devices as a phase-locking mechanism that induces localwave-form generation or oscillation, and may be of a transistor nature.Preferred interconnections or cross-couplings are by way of means thatgate energy to and from voltage supplies alternatively or additionallyto passing energy directly between the signal energy contra-flows in thesignal paths concerned. Inverter type interconnection circuits canafford both switching and amplification actions, say advantageouslyeffective to supply one direction of signal energy flow while absorbingreverse components in a laser-like action.

The endless electromagnetically continuous inverting signal paths ofapplication Ser. No. 09/529,076 and their reactive interconnections ofcomponent conductive elements combine to afford integration of signalwave-form generation and distribution, advantageously (but notnecessarily) of inherently fast rise/fall nature providing a remarkablegood substantially “square” wave-form even at very high plural-GigaHertzeffective frequencies. The contra-flow implementation of this inventioncan be in conjunction with use of an external exciter to launch thetraveling waves.

Another inventive feature inherent in application Ser. No. 09/529,076and now further developed and generalized in this continuation-in-partapplication concerns rotation locking as such. Application Ser. No.09/529,076 achieves rotation directionality in conjunction with energyconservation of its recirculatory signal energy flows combined withsignal generation, as specifically afforded by its endlesselectromagnetically continuous inverting signal paths.

However, signal rotation directionality can be achieved and maintainedby plural application of separately provided timing signals atprescribed positions spaced along a signal path for travelingelectromagnetic waves that is typically endless but need not be of anature applying signal inversion.

Typically, in implementing this inventive feature, three phases of inputtiming signals are connected at an endless non-inverting signal path atpositions appropriate to their phases.

A signal path comprising dual parallel conductive components, typicallytraces, can, as for the endless inverting signal paths of applicationSer. No. 09/529,076 and the above contra-flow inventive feature, providedifferential signal wave-forms at correlated take-off positions alongthe path. To this end, the plural phase input timing signals aresupplied with phase inversion to the two conductive components/traces,respectively at correlation positions along the path. Bipolar such inputtiming signals can be effective to make bipolar said wave-formsavailable round endless signal paths.

A signal path comprising a single conductive trace could, of course,provide a single-ended rather than differential signal path wave-form.

Another distinguishing feature is shared with the endlesselectromagnetically continuous inverting signal paths of applicationSer. No. 09/529,076 and the contra-flow unidirectional signal paths andthe plural-phase input signal provisions of this application. This otherdistinguishing feature is the use of signal paths of a transmission-linenature in association with non-linear active circuit elements. Suchnon-linear active provision can assist in sustaining signal flow energy,whether at application of plural phases to positions spaced along anendless electromagnetically continuous signal path or incross-connections between phase correlated positions along conductivesignal path components, as for dual conductive components/traces ofsignal paths as such in either of the rotationally endlesselectromagnetic continuity context of application Ser. No. 09/529,076 orin the contra-flow context of the first above feature hereof, or in theinterconnections between unidirectional signal paths of that featurehereof.

The cross-connections between dual conductive components/traces ofpreferred transmission line signal paths serve to continuously refresh atraveling voltage transition, preferably as very sharp voltagetransition as required for rise and fall of highly square wave-formsdesirable for clocking purposes. The interval between two opposite suchtransitions, or the actual or effective inversions of a singletransition, set the half-cycle time of a resulting wave-form that can beof bipolar nature.

Moreover, such transition regenerative action, along with take-off inthe manner of tapping into a passing wave-form, will be conservative ofpower requirements as a feature of timing signal distribution. This isparticularly advantageous for a large number of areally distributedsignal path provisions as can serve a large area semiconductorintegrated circuit, say of very large scale (VLSI) type. The abovecontra-flow embodiment is, of course, as in-principle readily arrayableas the endless electromagnetically continuous inverting signal pathswith hard-wired interconnects of application Ser. No. 09/529,076, withwhich it has high effective equivalence in terms of timing signaltake-off, though increasing the strength of its ordinaryinterconnections or cross-couplings would necessarily to some extentsacrifice chip area and increase power requirements.

Whilst the capability of the emphasized subject matter of applicationSer. No. 09/529,076 to operate at very high plural-GHz rates is sharedby the above one and further inventive features now emphasized herein,i.e. as inherently common to all three, the absence of integratedwave-form generation represented by requirement for a separatelygenerated input timing signal in the subject matter introduced hereincould be seen as leaving the matter of high-speed for such separatetiming signal provision unaddressed. However, there are and almostcertainly always will be, special technologies and fabrications that canprovide such a high speed source, albeit especially likely much moreexpensively and with less technologically coherence compared with theemphasized subject matter of application Ser. No. 09/529,076. However,the additional subject matter hereof does have application capability insuch manner.

It is feasible for the requisite stable high-frequency phase-reliableseparate timing signal source to be of the nature emphasized inapplication Ser. No. 09/529,076, then to have technical coherence withthe additional subject matter hereof, which is applicable asextension(s) of whatever provision is made by way of the emphasizedsubject matter of application Ser. No. 09/529,076.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is an outline diagram for one transmission-line structure hereof;

FIG. 2 shows a Moebius strip;

FIG. 3 is an outline circuit diagram for a traveling wave oscillator;

FIG. 4 is another outline circuit diagram for a traveling waveoscillator;

FIGS. 5 a and 5 b are equivalent circuits for distributed electricalmodels of a portion of a transmission-line of FIGS. 1–4;

FIG. 6 a shows related idealized graphs for respective differentialoutput waveforms;

FIG. 6 b illustrates relationship between propagation delay, electricallength and physical length of a transmission-line of FIGS. 1–4;

FIGS. 7(i)–7(ix) are idealized graphs illustrating the phase of relatedsignal waveforms;

FIGS. 8 a, 8 b illustrate instantaneous phasing of one waveform in atransmission-line oscillator;

FIG. 9 is a cross sectional view of part of a transmission-line on anIC;

FIGS. 10 a and 10 b are outline circuit and idealized graphs for astanding wave version;

FIG. 11 is a scrap outline of a transmission-line with invertingtransformer;

FIG. 12 shows a pair of back-to-back inverters connected across part ofa transmission-line;

FIGS. 13 a and 13 b are outline and equivalent circuit diagrams of CMOSback-to-back inverters;

FIG. 14 a details capacitive elements of a transmission-line togetherwith CMOS transistors;

FIG. 14 b shows an equivalent circuit diagram for FIG. 14 a;

FIG. 15 shows capacitive stub connections to a transmission-line;

FIG. 16 shows one connection for self-synchronizing transmission-lineoscillators;

FIGS. 17 a–17 c show other connections for self-synchronizingtransmission-line oscillators;

FIG. 18 is a diagrammatic equivalent representation for FIG. 17 a;

FIGS. 19 a and 19 b show connection of four transmission-lineoscillators;

FIGS. 20 and 21 show magnetically coupled self-synchronizedtransmission-line oscillators;

FIG. 22 shows three magnetically couple self-synchronizedtransmission-line oscillators;

FIG. 23 shows connection of self-synchronizing transmission-linesoscillators of different frequencies;

FIG. 24 shows an example of a clock distribution network for amonolithic IC;

FIG. 25 shows 3D implementation for timing systems hereof;

FIGS. 26 a and 26 b show examples of dual phase tap-off points;

FIG. 27 shows three concentrically arranged transmission-lineoscillators;

FIGS. 28 a and 28 b show a transmission-line having a cross-loopconnection;

FIG. 29 a shows a transmission-line configuration for four-phasesignals;

FIG. 29 b shows idealized resulting four-phase signal waveforms;

FIG. 30 shows an open-ended transmission-line connection;

FIG. 31 concerns coordinating frequency and phase for two IC's;

FIG. 32 shows digitally selectable shunt capacitors of Mosfet type;

FIG. 33 shows capacitive loading and routing data and/or power across atransmission line;

FIG. 34 is an outline circuit diagram for timing signal distributionusing transmission lines with contra-flow action with synchronization;

FIG. 35 a is an outline circuit diagram for timing signal availablewithout oscillation;

FIG. 35 b shows drive waveforms for the circuit of FIG. 35 a.

DETAILED DESCRIPTION FOR ILLUSTRATED EMBODIMENTS

Referring first to FIGS. 1–33, FIG. 1 shows a transmission-line 15 thatis neither terminated nor open-ended, nor even un-terminated as suchterm might be understood hitherto. As un-terminated, at least for use asin FIGS. 1–33, such transmission lines are seen as constituting astructural aspect of invention, including by reason of affording asignal path exhibiting endless electromagnetic continuity.

The transmission-line 15 of FIG. 1 is physically endless, specificallycomprising a single continuous “originating” conductor formation 17shown forming two appropriately spaced generally parallel traces asloops 15 a, 15 b with a cross-over at 19 that does not involve any localelectrical connection of the conductor 17. As shown, the length of theoriginating conductor 17 (taken as S), corresponds to two ‘laps’ of thetransmission-line 15 as defined between the spaced loop traces 15 a, 15b and through the cross-over 19.

This structure of the transmission-line 15 has a planar equivalence to aMoebius strip, see FIG. 2, where an endless strip with a single twistthrough 180.degree. has the remarkable topology of effectivelyconverting a two-sided and two-edged, but twisted and ends-joined,originating strip to have only one side and one edge, see arrowsendlessly tracking the centre line of the strip. From any position alongthe strip, return will be with originally left- and right-hand edgesreversed, inverted or transposed. The same would be true for any oddnumber of such twists along the length of the strip. Such a strip ofconductive material would perform as required for signal paths of firstpreferred embodiments of this invention, and constitutes anotherstructural aspect of invention. A flexible substrate would allowimplementing a true Moebius strip transmission-line structure, i.e. withgraduality of twist that could be advantageous compared with planarequivalent cross-over 19. A flexible printed circuit board so formed andwith its ICs mounted is seen as a feasible proposition.

FIG. 3 is a circuit diagram for a pulse generator, actually anoscillator, using the transmission-line 15 of FIG. 1, specificallyfurther having plural spaced regenerative active means conveniently asbi-directional inverting switching/amplifying circuitry 21 connectedbetween the conductive loop traces 15 a, 15 b. The circuitry 21 isfurther illustrated in this particular embodiment as comprising twoinverters 23 a, 23 b that are connected back-to-back. Alternativesregenerative means that rely on negative resistance, negativecapacitance or are otherwise suitably non-linear, and regenerative (suchas Gunn diodes) or are of transmission-line nature. It is preferred thatthe circuitry 21 is plural and distributed along the transmission-line15, further preferably evenly, or substantially evenly; also in largenumbers say up to 100 or more, further preferably as many and each assmall as reasonably practical.

Inverters 23 a, 23 b of each switching amplifier 21 will have the usualoperative connections to relatively positive and negative supply rails,usually V+ and GND, respectively. Respective input/output terminals ofeach circuit 21 are shown connected to the transmission-line 15 betweenthe loops 15 a, 15 b at substantially maximum spacing apart along theeffectively single conductor 17, thus each at substantially halfwayaround the transmission-line 15 relative to the other.

FIG. 4 is another circuit diagram for an oscillator using atransmission-line structure hereof, but with three crossovers 19 a, 19 band 19 c, thus the same Moebius strip-likereversing/inverting/transposing property as applies in FIG. 3.

The rectangular and circular shapes shown for the transmission-line 15are for convenience of illustration. They can be any shape, includinggeometrically irregular, so long as they have a length appropriate tothe desired operating frequency, i.e. so that a signal leaving anamplifier 21 arrives back inverted after a full ‘lap’ of thetransmission-line 15, i.e. effectively the spacing between the loops 15a,b plus the crossover 19, traversed in a time Tp effectively defining apulse width or half-cycle oscillation time of the operating frequency.

Advantages of evenly distributing the amplifiers 21 along thetransmission-line 15 are twofold. Firstly, spreading stray capacitanceeffectively lumped at associated amplifiers 21 for better and easierabsorbing into the transmission-line characteristic impedance Zo thusreducing and signal reflection effects and improving poor waveshapedefinition. Secondly, the signal amplitude determined by the supplyvoltages V+ and GND will be more substantially constant over the entiretransmission-line 15 better to compensate for losses associated with thetransmission-lines dielectric and conductor materials. A continuousclosed-loop transmission-line 15 with regenerative switching means 21substantially evenly distributed and connected can closely resemble asubstantially uniform structure that appears the same at any point. Agood rule is for elementary capacitance and inductance (Ce and Le)associated with each regenerative switching means and forming a resonantshunt tank LC circuit to have a resonant frequency of1/[2*pi*SQRT(Le/Ce)] that is greater than the self-sustainingoscillating frequency F (F3, F5 etc.) of the transmission-line 15.

FIG. 5 a is a distributed electrical equivalent circuit or model of aportion of a transmission-line 15 hereof. It shows alternate distributedresistive (R) and inductive (L) elements connected in series, i.e.R.sub.0 connected in series with L.sub.1 in turn connected in serieswith R.sub.2 and so on for a portion of loop 15 a, and registeringL.sub.0 connected in series with R.sub.1 in turn connected in serieswith L.sub.2 and so on for the adjacent portion of loop 15 b; anddistributed capacitive elements C.sub.0 and C.sub.1 shown connected inparallel across the transmission-line 15 thus to the loops 15 a and 15 bbetween the resistive/inductive elements R.sub.0/L.sub.1 and theinductive/resistive elements L.sub.0/R.sub.1, respectively for C.sub.0,and between the inductive/resistive elements L.sub.1/R.sub.2 and theresistive/inductive elements R.sub.1/L.sub.2, respectively for C.sub.1:where the identities R0=R1=R2, L1=L2=L3 and CO=C1 substantially hold andthe illustrated distributed RLC model extends over the whole length ofthe transmission-line 15. Although not shown, there will actually be aparasitic resistive element in parallel with each capacitive element C,specifically its dielectric material.

FIG. 5 b is a further simplified alternative distributed electricalequivalent circuit or model that ignores resistance, see replacement ofthose of FIG. 5 a by further distribution of inductive elements inseries at half (L/2) their value (L) in FIG. 5 a. This model is usefulfor understanding basic principles of operation of transmission-linesembodying the invention.

During a ‘start-up’ phase, i.e. after power is first applied to theamplifiers 21, oscillation will get initiated from amplification ofinherent noise within the amplifiers 21, thus begin substantiallychaotically though it will quickly settle to oscillation at afundamental frequency F, typically within nano-seconds. For eachamplifier 21, respective signals from its inverters 23 a and 23 b arriveback inverted after experiencing a propagation delay Tp around thetransmission-line 15. This propagation delay Tp is a function of theinductive and capacitive parameters of the transmission-line 15; which,as expressed in henrys per meter (L) and in farads per meter (C) toinclude all capacitive loading of the transmission-line, lead to acharacteristic impedance Zo=SQR (L/C) and a line traverse or propagationor phase velocity Pv=1/SQRT(L*C). Reinforcement, i.e. selectiveamplification, of those frequencies for which the delay Tp is an integersub-divisor of a half-cycle time gives rise to the dominant lowestfrequency, i.e. the fundamental frequency F=1/(2.multidot.Tp), for whichthe sub-divisor condition is satisfied. All other integer multiples ofthis frequency also satisfy this sub-divisor condition, but gain of theamplifiers 21 ‘falls off’, i.e. decreases, for higher frequencies, sothe transmission-line 15 will quickly settle to fundamental oscillationat the frequency F.

The transmission-line 15 has endless electromagnetic continuity, which,along with fast switching times of preferred transistors in theinverters 23 a and 23 b, leads to a strongly square wave-form containingodd harmonics of the fundamental frequency F in effectively reinforcedoscillation. At the fundamental oscillating frequency F, including theodd harmonic frequencies, the terminals of the amplifiers 21 appearsubstantially unloaded, due to the transmission-line 15 being‘closed-loop’ without any form of termination, which results verydesirably in low power dissipation and low drive requirements. Theinductance and capacitance per unit length of the transmission-line 15can be altered independently, as can also be desirable and advantageous.

FIG. 6 a shows idealized waveforms for a switching amplifier 21 withinverters 23 a and 23 b Component oscillation waveforms .PHI.1, .PHI.2appear at the input/output terminals of that amplifier 21 shortly afterthe ‘start-up’ phase, and continue during normal operation. Thesewaveforms .PHI.1 and .PHI.2 are substantially square and differential,i.e. two-phase inverse in being 180 degrees out-of-phase Thesedifferential waveforms .PHI.1 and .PHI.2 cross substantially at themid-point (V+/2) of the maximum signal amplitude (V+). This mid point(V+/2) can be considered as a ‘null’ point since the instant that boththe waveforms .PHI.1 and .PHI.2 are at the same potential, there is nodisplacement current flow present in nor any differential voltagebetween the conductive loop traces 15 a and 15 b. For the preferredrecirculating traveling wave aspect of this invention, this null pointeffectively sweeps round the transmission line 15 with very fast riseand fall times and a very ‘clean’ square-wave form definition. This nullpoint is also effectively a reference voltage for opposite excursions ofa full cycle bipolar clock signal.

For the transmission-line 15, it is convenient to consider complete lapsas traversed by a traveling wave, and also total length S of theoriginating conductive trace 17, both in terms of ‘electrical length’.FIG. 6 b shows relationships between the propagation delay or traversetime (Tp), electrical length in degrees, and physical length (S) oforiginating conductive line/trace 17. For each of the out-of-phasewaveforms .PHI.1 and .PHI.2, and as seen by a traveling wave repeatedlytraversing the transmission-line 15, each substantially square waveexcursion corresponds to one complete lap, i.e. one traverse time Tp,and successive opposite wave excursions require two consecutive laps,i.e. two traverse times (2.times.Tp). One lap of the transmission-line15 thus has an ‘electrical length’ of 180 degrees, and two laps arerequired for a full 0.degree.–360.degree. bipolar signal cycle, i.e.corresponding to the full lengths of the originating conductor 17.

By way of example, an electrical length of 180.degree. corresponding toone lap and ½ wavelength at 1 GHz could be formed from a 45 mmtransmission-line having a phase velocity (Pv) that is 30% that of thespeed of light (c), i.e. Pv=0.3*c, or 4.5 mm where Pv=0.03*c, or 166 mmin free space, i.e. where Pv=1*c.

FIGS. 7( i)–7(ix) show waveforms .PHI.1, .PHI.2 through a full cycle tostart of the next cycle, specifically at eight equal electrical-lengthspacings of 45 degrees between sample positions along the conductor lineor trace 17. Phase labeling are relative to FIG. 7( i) which can beanywhere along the trace 17, i.e. twice round the transmission line 15,as such, and 0/360-degrees for rise/fall of the .PHI.1, .PHI.2 waveforms15 is arbitrarily marked. Taking FIG. 7(i) as time t0, FIG. 7(ii) showsthe waveforms .PHI.1, .PHI.2 at time t0+(0.25 Tp) after one-eighth(0.125 S) traverse of total length S of the line 17, thus traverse ofone-quarter of the transmission line 15, and 45-degrees of electricallength. Times t0+(0.5 Tp), t0+(0.75 Tp), t0+(0.75 Tp) . . . t0+(2 Tp);traverses 0.25 S, 0.375 S, 0.5 S . . . 1.0 S and 90, 135, 180 . . .360-degrees should readily be seen self-evidently to apply to FIGS.7(iii)–(ix), respectively.

FIGS. 8 a and 8 b show snap-shots of excursion polarity (shown circled),displacement current flow (shown by light on-trace arrows), andinstantaneous phasing from an arbitrary 0/360-degree position on theelectromagnetically endless transmission line 15 covering two lapsthereof (thus the full length the continuous originating conductor 17).Only one differential traveling electromagnetic (EM) waveform (say.PHI.1) of FIG. 7 is shown, but for rotation propagation around thetransmission-line 15 in either of opposite directions, i.e. clockwise orcounter-clockwise. The other waveform (Φ2) will, of course be180.degree. out of phase with the illustrated waveform (Φ1). The actualdirection of rotation of the EM wave will be given by Poyntings' vector,i.e. the cross product of the electric and magnetic vectors. Thecrossover region 19 produces no significant perturbation of the signalsΦ1 or Φ2 as the EM wave traverses this region 19. In effect, the fastrise/fall transitions travel round the transmission-line at phasevelocity Pv, the switching amplifiers 21 serving to amplify thetransitions during first switching between supply voltage levels.

The phases of the waveforms Φ1 and Φ2 can, for a transmission-line 15hereof, be accurately determined from any arbitrary reference point onthe transmission-line 15, thus have strong coherence and stability ofphasing.

Suitable (indeed preferred in relation to present IC manufacturingtechnology and practice) switching amplifiers 21 for bidirectionaloperation are based on back-to-back Mosfet inverters 23 a,b, for whichup to well over 1,000 switching inverting amplifier pairs could beprovided along typical lengths of transmission-line structures hereof.

The bidirectional inverting action of the switching amplifiers 21 is ofsynchronous rectification nature. The rise and fall times of thewaveforms Φ1 and Φ2 are very fast indeed compared with hithertoconventional timing signals, being based on electron-transit-time ofpreferred Mosfet transistors of the inverters 23 a,b. Moreover,reinforcement is related to the transmission-line 15 having lowerimpedance than any ‘on’ transistor in inverters of preferredbidirectional switching amplifiers 21, though total paralleled isusefully of the same order. Switching of such inverters means that eachamplifier 21 contributes to the resulting wave polarity by way of asmall energy pulse which, by symmetry, must propagate in bothdirections, the forwardly directed EM wave pulse thus contributing asdesired. The reverse EM wave pulse that travels back to the previouslyswitched amplifier 21 is of the same polarity as already exists there,thus reinforces the preexisting switched state. Ohmic paths betweenpower supply rails and the transmission line 15 through ‘on’ transistorsof the preferred inverters of amplifiers 21 ensure that energy of suchreverse EM wave pulses is absorbed into those power supply rails V+,GND,i.e. there is useful power conservation.

It should be appreciated that implementation could be by other thanCMOS, e.g. by using N-channel pull-ups, P-channel pull-downs, bipolartransistors, negative resistance devices such as Gunn diodes, Mesfet,etc.

Regarding the transmission-lines 15 as such, a suitable medium readilyapplicable to ICs and PCBs and interconnects generally is as commonlyreferred to as microstrip or coplanar waveguide or stripline, and wellknown to be formable lithographically, i.e. by patterning of resists andetching. Practical dielectrics for an on-IC transmission-line includesilicon dioxide (SiO.sub.2) often referred to as field oxide,inter-metal dielectrics, and substrate dielectrics (which can be used atleast for semi-insulating structures, e.g. of silicon-on-insulatortype).

FIG. 9 is a cross-section through a portion of one exemplary on-ICtransmission-line formation comprising three metal layers 56, 58 and 60and two dielectric layers 62 and 64. Middle metal layer 58 isillustrated as comprising the two transmission-line loop conductivetraces 15 a and 15 b that are at least nominally parallel. Upper metallayer 60 could be used as an AC ‘ground’ plane and could be connected tothe positive supply voltage V+, lower metal 56 being a ‘ground’ planethat could be connected to the negative supply voltage GND. Thedielectric layers 62 and 64 between the metal transmission-line tracesat 58 and ‘ground’ planes 56 and 58 are typically formed using silicondioxide (SiO2). The full illustrated structure is seen as preferable,though maybe not essential in practice, i.e. as to inclusion of eitheror both of the ‘ground’ planes and the dielectric layers 62, 64. Thephysical spacing 66 between the conductive traces 15 a, 15 b affects thedifferential and common modes of signal propagation, which shouldpreferably have equal, or substantially equal, velocities in order toachieve minimum dispersion of the electromagnetic field from the spacing66. Screening properties improve with use of ‘ground planes’, as doesthe ability for the structure to drive non-symmetrical, i.e. unbalanced,loads applied to the conductive traces 15 a, 15 b.

Inter-metal dielectric layers on a typical IC CMOS process are thin,typically about 0.7.μm, so microstrip transmission-line features withlow signal losses must have a low characteristic impedance Zo (ashitherto for un-terminated, partially terminated or series terminatedlines acting to reduce signal reflections to a manageable level).Self-sustaining, non-terminated, closed-loop transmission-lines 15hereof inherently have very low power consumption for maintainedtraveling EM wave oscillation as the dielectric and conductor losses tobe overcome are typically low. From FIG. 5 b, it will be appreciatedthat, if there were no resistive losses associated with thetransmission-line 15 and amplifiers 21, the transmission-line 15 wouldrequire no more energy than required initially to ‘charge-up’ thetransmission-lines inductive Le and capacitive Ce elements. The EM wavewould continually travel around the transmission-line with all energy inthe transmission-line 15 simply transferred, or recycled between itselectric and magnetic fields, thus capacitive Ce and inductive Leelements. Whilst there must be some resistive losses associated with thetransmission-line 15 and amplifiers 21, see transmission-line resistiveelements R.sub.0-R.sub.2 in FIG. 5 a, the resistance is typically lowand associated resistive losses will be also low. There is no penaltyherein from for using low-impedance transmission-lines 15, evenadvantage from being less affected by capacitive loading, thus resultingin ‘stiffer’ drive to logic gates.

A crossover 19 can be implemented on an IC using ‘vias’ between themetal layers, preferably with each via only a small fraction of totallength S of the transmission-line 15.

A variant is available where a transmission-line 15 hereof has only oneamplifier 21 connected to the transmission-line, and the EM wave nolonger travels around the transmission-line 15 so that a standing waveoscillation results, see FIG. 10 a for single amplifier 21 and FIG. 10 bfor differential waveforms. Such amplifier should not extend over morethan approximately 50 of the electrical length of the transmission-line15. If the single amplifier 21 never goes fully ‘on’ or ‘off’ a standingsine wave oscillation will result in the transmission-line 15, whichwill have varying amplitude with the same phases at the same positionsincluding two stationary, two ‘null regions.

It follows that traveling wave operation will be available using a fewspaced or just one lengthy CMOS bidirectional inverter formation, thoughplural small inverters will produce smoother faster results. Offsettingformations of the amplifiers 21, even just its input/output terminals,can predispose a traveling EM wave to one direction of transmission-linetraversal, as could specific starter circuit such as based on forcingfirst and slightly later second pulses onto the transmission-line atdifferent positions, or incorporation of some known microwavedirectional coupler.

Inverting transmission-line transformers can be used instead of thecrossovers (19) and still yield a transmission line having endlesselectromagnetic continuity, see FIG. 11 for scrap detail at 21T.

FIG. 12 shows a pair of back-to-back inverters 23 a, 23 b with supplyline connectors and indications of distributed inductive (L/2) andcapacitive (C) elements of a transmission-line as per FIG. 5 b. FIG. 13a shows N-channel and P-channel Mosfet implementation of theback-to-back inverters 14 a and 14 b, see out of NMOS and PMOStransistors.

FIG. 13 b shows an equivalent circuit diagram for NMOS (N1, N2) and PMOS(P1, P2) transistors, together with their parasitic capacitances. Thegate terminals of transistors P1 and N1 are connected to the conductivetrace 15 a and to the drain terminals of transistors P2 and N2.Similarly, the gate terminals of transistors P2 and N2 are connected tothe conductive trace 15 b and to the drain terminals of transistors P2and N2. The PMOS gate-source capacitances CgsP1 and CgsP2, the PMOSgate-drain capacitances CgdP1 and CgdP2, and the PMOS drain-source andsubstrate capacitances CdbP1 and CdbP2, also the NMOS gate-sourcecapacitances CgsN1 and CgsN2, the NMOS gate-drain capacitances CgdN1 andCgdN2, and the NMOS drain-source and substrate capacitances CdbN1 andCdbN2 are effectively absorbed into the characteristic impedance Zo ofthe transmission-line, so have much less effect upon transit times ofthe individual NMOS and PMOS transistors. The rise and fall times of thewaveforms .PHI.1 and .PHI.2 are thus much faster than for priorcircuits.

For clarity FIGS. 12–14 omit related resistive (R) elements. FIG. 14 ashows only the capacitive elements (as per FIGS. 12 and 13 b) of thetransmission-line 15 together with those of the N/PMOS transistors. FIG.14 b illustrates another equivalent circuit diagram for FIG. 14 aincluding the transmission-line distributed inductive (L/2) elements andthe effective capacitance Ceff given by:Ceff=C+CgdN+CgdP+[(CgsN+CdbN+CgsP+CdbP)/4];

Where:

CgdN=CgdN1+CgdN2;

CgdP=CgdP1+CgdP2;

CgsN=CgsN1+CgsN2;

CdbN=CdbN1+CdbN2;

CgsP=CgsP1+CgsP2; and

CdbP=CdbP1+CdbP2.

Capacitance loading due to gate, drain, source and substrate junctioncapacitances are preferably distributed as mentioned previously.

An advantage of having a differential- and common-mode,transmission-line, is that ‘parasitic’ capacitances inherent withinmosfet transistors can be absorbed into the transmission-line impedanceZo, as illustrated in FIGS. 14 a and 14 b, and can therefore be used forenergy transfer and storage. The gate-source capacitances (Cgs) of theNMOS and PMOS transistors appear between the signal conductor traces 15a, 15 b and their respective supply voltage rails and can be compensatedfor by removing the appropriate amount of respective capacitance fromconnections of the transmission-line 15 to the supply voltage rails, sayby thinning the conductor traces 15 a, 15 b by an appropriate amount.The gate-drain capacitance (Cgd) of the NMOS and PMOS transistors appearbetween the conductive traces 15 a and 15 b and can be compensated forby proportionally increasing the spacing 66 between the conductivetraces 15 a, 15 b at connections to the NMOS and PMOS transistors of theinverters 23 a/b.

By way of a non-restrictive example, on a 0.35 micron CMOS process, ausable 5 GHz non-overlapping clock signal should result withtransmission-line loop length (S/2) of 9 mm for a phase velocity of 30%of speed-of-light, as determined by capacitive shunt loadingdistribution and dielectric constants, the total length (S), of theconductor 17 thus being 18 mm.

The substrate junction capacitances (Cdb) of the NMOS and PMOStransistor could be dramatically reduced by using semi-insulating orsilicon-on-insulator type process technologies.

There is a continuous DC path that directly connects the terminals ofeach of the amplifiers 21, i.e. the respective input/output terminals ofeach and all of the inverters 23 a, 23 b, but this path is characterizedby having no stable DC operating point. This DC instability isadvantageous in relation to the regenerative action of each of therespective amplifiers 21.sub.1–21.sub.4 and their positive feedbackaction.

Transmission-lines 15 hereof can be routed around functional logicblocks as closed-loops that are ‘tapped into’ to get ‘local’ clocksignals. CMOS inverters can be used as ‘tap amplifiers’ in a capacitive‘stub’ to the transmission-line 15, which can be ‘resonated out’ byremoving an equivalent amount of ‘local’ capacitance from thetransmission-lines, say by local thinning of conductor traces (15 a/15b) as above. Capacitive ‘clock taps’ can be spread substantially evenlyalong a transmission-line 15 hereof having due regard as a matter ofdesign to their spacings, which, if less than the wavelength of theoscillating signal, will tend to slow the propagation of the EM wave andlower the characteristic impedance Zo of the transmission-line (15), butwill still result in good signal transmission characteristics.

Within functional logic blocks that are small relative to clock signalwavelength, un-terminated interconnects work adequately for localclocking with phase coherence, see FIG. 15. For clarity, the pairs ofconnections to the transmission-line 15 are shown slightly offset,though they would typically be opposite each other in practice.Alternative tap-off provisions include light bidirectional of passiveresistive, inductive or transmission-line nature, or unidirectional orinverting connections, including much as for what will now be describedfor interconnecting transmission-lines 15 themselves.

Plural oscillators and transmission-lines 15 can readily be operativelyconnected or coupled together in an also inventive manner, includingsynchronizing with each other both in terms of phase and frequencyprovided that any nominal frequency mismatch is not too great.Resistive, capacitive, inductive or correct length directtransmission-line connections/couplings, or any combinations thereof,can make good bidirectional signal interconnections. Signal connectionor coupling between transmission-lines can also be achieved using knowncoupling techniques as used for microwave micro-strip circuits,generally involving sharing of magnetic and/or electrical flux betweenadjacent transmission lines. Unidirectional connections can also beadvantageous. Connectors and couplings hereof are capable of maintainingsynchronicity and coherency of plural transmission-line oscillatorsthroughout a large system, whether within ICs or between IC's say onprinted circuit boards (PCBs).

Connection/coupling of two or more transmission-lines andcross-connection rules are similar to Kirchoff's current law but basedon the energy going into a junction, i.e. a connection or coupling, ofany number of the transmission-lines being equal to the energy comingout of the same junction, i.e. there is no energy accumulation at thejunction. When the supply voltage V+ is constant, the rule is, ofcourse, precisely Kirchoff's current law. By way of a practical example,if there is a junction common to three transmission-lines, the simplest,but not the only, solution is that one of the transmission-lines hashalf the characteristic impedance of the other two transmission-lines.Where there are any even number of coupled transmission-lines, theirrespective characteristic impedances can all be equal. However, thereare an infinite number of combinations of impedances which will satisfyKirchoff's current law. The cross-connection rule, within atransmission-line, is the same as the rules for coupling two or moretransmission-lines described above.

There will be high quality differential signal waveforms .PHI.1 and.PHI.2, in terms of phase and amplitude, at all points around atransmission-line network 15 when the following criteria are met:

(i) the transmission-lines have substantially matching electricallengths

(ii) above Kirchoff-like power rules are satisfied

(iii) there is phase inversion.

There are, of course, an infinite number of coupled network designs andsupply voltages that will fulfill the above three criteria, such as forexample: short sections of slow, low impedance transmission-lines thatare coupled to long fast, high impedance transmission-lines; and one-and/or three-dimensional structures etc. However, for the bestwave-shapes and lowest parasitic power losses, the phase velocities ofthe common-mode and the differential-mode, i.e. even and odd modes,should be substantially the same. The same, or substantially the same,phase velocities can be designed into a system by varying thecapacitances of the transmission-lines.

The supply voltage V+ does not have to be constant throughout a system,provided that above Kirchoff-like power/impedance relationships aremaintained and result in an inherent voltage transformation system that,when combined with the inherent synchronous rectification of theinverters 23 a and 23 b, allows different parts of the system to operateat different supply voltages, and power to be passed bi-directionallybetween such different parts of the system.

FIG. 16 shows two substantially identical transmission-line oscillatorshereof that are operatively connected such that they are substantiallyself-synchronizing with respect to frequency and phase. Thetransmission-lines 15.sub.1 and 15.sub.2 are shown ‘siamesed’ with thecommon part of their loop conductive traces meeting above Kirchoff-likepower/impedance rule by reason of its impedance being half theimpedances (20) of the remainders of the transmission-lines 15.sub.1 and15.sub.2, because the common parts carry rotating wave energy of both ofthe two transmission-lines 15.sub.1 and 15.sub.2. As noted above, theoriginating trace length S of a transmission-line is one factor indetermining the frequency of oscillation so transmission-lines 15.sub.1and 15.sub.2 using the same medium and of substantially identical lengthS will have substantially the same frequency of oscillation F and willbe substantially phase coherent. In FIG. 16, respective EM waves willtravel and re-circulate in opposite directions around thetransmission-lines 15.sub.1 and 15.sub.2, see marked arrows 1L, 2L (orboth opposite), in a manner analogous to cog wheels. Such siamesingconnection of transmission-lines can readily be extended sequentially toany number of such ‘cogged’ transmission-line oscillators.

FIG. 17 a shows another example of two substantially identicaltransmission-line oscillators with their transmission lines 15, and 152operatively connected to be substantially self-synchronizing infrequency and phase by direct connections at two discrete positions 40and 42. FIG. 17 b shows such direct connections via passive elements 44,46 that could be resistive, capacitive or inductive or any viablecombination thereof. FIG. 17 c shows such direct connections viaunidirectional means 48 that can be two inverters 50.sub.1 and 50.sub.2.The unidirectional means 48 ensures that there is no coupling or signalreflection from one of the transmission-lines (15.sub.2) back into theother (15.sub.1), i.e. only the other way about. Directions of travel ofre-circulating EM waves are again indicated by arrows 1L, 2L that aresolid but arbitrary for transmission-line oscillator 15.sub.1 and dashedfor 15.sub.2 in accordance with expectations as to a ‘parallel’-coupledpair of transmission-lines yielding contra-directional traveling waves.FIG. 18 is a convenient simplified representation of the twoself-synchronized transmission-line oscillators of FIG. 17 a, andsimilar representations will be used in following Figures.

FIG. 19 a shows four self-synchronized transmission-line oscillators15.sub.1–15.sub.4 connected together basically as for FIGS. 17 a-17 c,but so as further to afford a central fifth effective transmission-linetiming signal source of this invention affording a re-circulatorytraveling EM wave according to indicated EM wave lapping directions1L–4L of the four transmission-line oscillators 15.sub.1–15.sub.4. Asshown the central fifth transmission-line oscillator physicallycomprises parts of each of the other four, and has a lapping direction5L that is opposite to theirs, specifically clockwise forcounter-clockwise 1L–4L. It will be appreciated that this way ofconnecting transmission-line oscillators together can also be extendedto any desired number and any desired variety of overall pattern tocover any desired area.

An alternative is shown in FIG. 19 b where the central fifthtransmission-line oscillator is not of re-circulating type, but isnonetheless useful and could be advantageous as to access to desiredphases of timing signals.

FIG. 20 shows two self-synchronizing oscillators with theirtransmission-lines 15.sub.1 and 15.sub.2 not physically connectedtogether, rather operatively coupled magnetically; for which purpose itcan be advantageous to use elongated transmission-lines to achieve moreand better magnetic coupling. FIG. 21 shows another example ofmagnetically coupled self-synchronizing oscillators withtransmission-lines 15.sub.1 and 15.sub.2 generally as for FIG. 20, butwith a coupling enhancing ferromagnetic strip 52 operatively placedbetween adjacent parts to be magnetically coupled.

FIG. 22 shows three self-synchronizing oscillators with theirtransmission-lines 15.sub.1, 15.sub.2 and 15.sub.3 magnetically coupledby a first ferrous strip 52 placed between transmission-lines 15.sub.1and 15.sub.2 and a second ferrous strip 54 placed betweentransmission-lines 15.sub.2 and 15.sub.3. As a source of oscillatingsignals, the transmission-line 15.sub.2 does not need any regenerativeprovisions 21 so long as enough energy for oscillation is magneticallycoupled from the other transmission-lines 15.sub.1 and 15.sub.3 that arecomplete with provisions 21. It is considered practical for thetransmission-line 15.sub.2 to be longer and circumscribe a larger areabut not to need or have regenerative provisions 21, nor a cross-over 19;and is then preferably an odd multiple (3 S, 5 S, 7 S etc) of the length(S) or at least the electrical length of at least one of thetransmission-lines 15.sub.1 and 15.sub.3. This, of course, has furtherimplications for self-synchronizing frequency- and phase-locking ofoscillators (say as using transmission-lines 15.sub.1 and 15.sub.3), ata considerable spacing apart.

Further alternatives include use of a dielectric material (notillustrated) that spans over and/or under the portions of the conductivetraces to be electromagnetically coupled.

It is feasible and practical to synchronize transmission-lineoscillators operating at different frequencies. In FIG. 23,transmission-lines of two self-synchronizing oscillators are ofdifferent electrical lengths. Specifically, using same transmission-linestructure/materials, first transmission-line 15, has a total conductivelength S for a fundamental oscillating frequency F=F1 and is operativelyconnected and synchronized to a second transmission-line 15.sub.2 havinga total conductive length that is one third of that of the firsttransmission-line 15.sub.1, i.e. S/3, thus an oscillating frequency of 3F. The dashed lines with arrows indicate the direction of rotation ofthe EM waves. Operative connection is as for FIGS. 17 a–c, though anyother technique could be used. Self-synchronizing is due toabove-mentioned presence in the highly square first transmission-linesignal of a strong third harmonic (3 F). Similar results are availablefor higher odd harmonics, i.e. at frequencies of 5 F, 7 F etc.

Preferred coupling between transmission-lines of oscillators operatingat such different odd harmonic related frequencies, is unidirectional sothat the naturally lower frequency line (15.sub.1) is not encouraged totry to synchronize to the naturally higher frequency line (15.sub.2).Any number of transmission-line oscillators of differentodd-harmonically related frequencies can be coupled together andsynchronized as for FIG. 23.

Re-circulatory transmission-line oscillators hereof can be used in andfor the generation and distribution of reference, i.e. clock, timingsignal(s) in and of a semiconductor integrated circuit (IC); and is alsoapplicable to a printed-circuit-board (PCB), e.g. as serving to mountand interconnect circuitry that may include plural ICs, or indeed, anyother suitable apparatus/system where timing reference signal(s) is/arerequired.

For ICs as such, simulations using the industry standard SPICEtechniques show potential for supplying clock signals of very highfrequencies indeed, up to several tens of GHz, depending upon the ICmanufacturing process employed and projections for their development.Generation and distribution can effectively be at, and service, allparts of an IC with predictable phases at and phase relationshipsbetween such parts, including as multiple clock signals that may havethe same or different frequencies. Moreover, principles of operation oftransmission-line oscillators hereof and their self-synchronizinginter-coupling extend or lead readily not only to reliable service oftiming signals to operational circuitry within any particular IC andbetween ICs, but further and it is believed also importantly andinventively to data transfer between ICs etc.

The entire transmission-line 15 structure and network involvingregenerative circuits 21 oscillates. The transmission-line 15 operatesun-terminated, i.e. the transmission-line forms a closed-loop. Thecharacteristic impedance Zo of the transmission-line is low and only‘top-up’ energy is required to maintain oscillation.

Impedance between the two conductor traces 15 a, 15 b is preferablyevenly distributed, thus well balanced, which helps achieve welldefined, differential signal waveforms (.PHI.1, .PHI.2). Coherentoscillation occurs when the signals .PHI.1, .PHI.2 on thetransmission-line 15 meet this 180.degree., or substantially a180.degree., phase shift requirement for all inverting amplifiers 21connected to the transmission-line 15 i.e. when all the amplifiers 21operate in a coordinated manner with known phase relationship betweenall points along the transmission-line 15. Signal energy is transmittedinto the transmission-line 15 both inductively and capacitively, i.e.magnetically and electrically, between the signal conductors 15 a, 15 bfor the differential-mode, also between each signal conductor and theground reference for the two individual common-mode (not present if theupper and lower ‘ground’ planes are absent, nor for connections viaunshielded twisted-pair cables).

CMOS inverters as non-linear, operative switching and amplifying circuitelements have low losses from cross-conduction current as normally lossytransistor gate ‘input’ and drain ‘output’ capacitances are absorbedinto the characteristic impedance Zo of the transmission-line 15, alongwith the transistor substrate capacitances, so power consumption is notsubject to the usual ½.C.V.sup.2.f formula.

It is quite often assumed that the power dissipation due to capacitivecharging and discharging of MOS transistor gates, for example, isunavoidable. However, the self sustaining oscillating nature of thetransmission-line 15 is able to ‘drive’ the transistor gate terminalswith low power loss. This is due to the fact that the required ‘drive’energy is alternating between the electrostatic field, i.e. thecapacitive field of the MOS gate capacitances, and the magnetic field,i.e. the inductive field elements of the transmission-line 15.Therefore, the energy contained within the transmission-line 15 is notbeing completely dissipated, it is in fact being recycled. Energy savingapplies to all operatively connected transistor gates of thetransmission-line 15.

It is envisaged that low loss efficiency of transmission-line oscillatorhereof could well be used to ‘clock’ ICs for many previously popularlogic systems that have since been overshadowed or abandoned asnon-viable options for reasons attributed to problems associated withclock skew, clock distribution, power consumption etc. Non-exhaustiveexamples of such logic arrangements include poly-phase logic and chargerecovery or adiabatic switching logic, such logic arrangements beingknown to those skilled in the art.

FIG. 24 shows a possible clock distribution network hereof as applied toa monolithic IC 68 (not to scale, as is other Figures hereof). The IC 68has a plural transmission-lines hereof shown as loops 1L–13L, of whichloops 1L–10L and 13L all have the same effective lengths (say as for Sabove) and oscillate at a frequency F, and loops 11L and 12L each haveshorter loop lengths (say as for S/3 above) and oscillate at a frequency3 F. Loops 1L–8L and 11L–13L are full transmission-line oscillatorcomplete with regenerative means, and loops 9L and 10L arise as parts offour of the former transmission-lines, namely 1L, 3L, 4L and 5L; 4L, 5L,6L and 8L respectively.

The transmission-line (15) of the loop 13L is elongated with a long sideclose to the edge (i.e. scribe line) of the IC 68, so that it ispossible to couple to another similarly set up separate monolithic ICfor inter-coupling by such as flip-chip technology for frequency andphase locking by such as magnetic coupling, as described above. Phaseand frequency locking of separate monolithic IC's can be very useful insuch as hybrid systems.

FIG. 25 indicates feasibility of a three-dimensional network ofinterconnected transmission line oscillators hereof for signaldistribution, specifically for a simple pyramidal arrangement, thoughany other structure could be serviced as desired, no matter how complexso long as interconnect rules hereof are met regarding electricallength, impedance matching, any phasing requirements for data transfer,etc.

ICs hereof can be designed to have whatever may be desired up to totalfrequency and phase locking, also phase coherence, including for andbetween two or more self-sustaining transmission-line oscillatorsgreatly to facilitate synchronous control and operation of dataprocessing activities at and between all the various logic andprocessing blocks associated with such IC.

FIG. 26 a shows an example of dual phase tap-off using a pair of CMOSinverters 70.sub.1 and 70.sub.2 connected to the transmission-lineconductive traces 15 a and 15 b respectively to provide local clock toand/or to be distributed about a logic block 72.sub.1. Whilst the logicblock 72.sub.1 is shown as being ‘enclosed’ within the transmission-line15 alternatives include it being outside any area enclosed by thetransmission-line 15, as for the logic block 72.sub.2 and its associatedinverters 70.sub.3, 70.sub.4, and/or it spanning the conductive traces15 a, 15 b of the transmission line 15. If desired, say for large logicblocks 72.sub.1 and/or 72.sub.2 plural pairs of inverters 70 can ‘tap’into the transmission-line 15, including for any desired phasing neededlocally in the logic block 72, see dashed line. Capability accurately toselect the phase of the oscillating clock signals .PHI.1, .PHI.2 allowscomplex pipeline logic and poly-phase logic (see FIG. 29 below) to beoperatively designed and controlled.

FIG. 26 b differs in that the logic blocks 71.sub.1, 72.sub.2 arereplaced by respective processing elements 73.sub.1, 73.sub.2, thoughthere could be more, and for which one or more transmission-lines can beused to clock one or more of the processing elements. Two or a greaterplurality of processing elements can operate independently and/ortogether, i.e. in parallel to achieve very fast and powerful dataprocessing ICs/systems.

FIG. 27 a shows concentrically arranged transmission-lines15.sub.1–15.sub.3 of progressively less physical lengths. However, eachof the three transmission-lines 15.sub.1-–5.sub.3 can be made so thatthey all oscillate at the same frequency, whether as a matter ofstructure or by respective velocities of the EM waves rotating aroundeach of the shorter transmission-lines 15.sub.2 and 15.sub.3 beingsuitably retarded by increasing their inductance and/or capacitance perunit length. Moreover, the transmission-lines 15.sub.1–15.sub.3 canoptionally have one or more operative connections 70 and 72 that willserve to synchronize the three transmission-lines 15.sub.1–15.sub.3. Theadvantages, apart from synchronicity, of having these connections 70, 72are that the transmission-lines 15.sub.1–15.sub.3 will or can

(i) act as a single multi-filament transmission-line;

(ii) have smaller conductive traces (15 a, 15 b);

(iii) cover a larger clocking area;

(iv) produce lower skin effect losses; and

(v) produce lower crosstalk and coupling.

FIG. 28 a shows a transmission-line having a cross-loop connectionbetween positions A, B, C and D, which comprises furthertransmission-line 15 c, 15 d that has, in this particular example, anelectrical length of 90.degree. to match spacing of the positions A, Band C, D. Other cross-connection electrical length could be chosen, thenoperatively connected at correspondingly different spacings of thepositions A, B and C, D. Cross-loop connections allow further tap-offpositions within area enclosed by the transmission-line 15. Thetransmission-line part 15 d is shown connected in parallel, betweenpoints A and C, and part of the transmission-line 15 represented by line74. Likewise, the transmission-line part 15 c is shown connected inparallel, between points B and D, with part of the transmission-line 15represented by line 76. The transmission-line parts 15 c, 15 d, 74 and76 will be satisfactory if they each have an impedance that is half thatassociated with the remainder of the transmission-line 15, as above. Thetransmission-lines 15 and 15 c,d will have operatively connectedamplifiers 21. FIG. 28 b shows the cross-loop connection 15 c,d and thepositions A, B, C and D set up relative to parts 78 and 80 of thetransmission-line 15, i.e. instead of parts 74 and 76, respectively; butwith Kirchoff-type rules applying again to result in parts 15 c, 15 d,78 and 80 each having an impedance of half that associated with theremainder of the transmission-line 15. Introduction of plural additionaltransmission-lines such as 15 c,d across a transmission-line 15 isfeasible as required.

FIG. 29 a shows one way to produce four-phase clock signals.Effectively, a transmission-line 15 makes a double traverse of itssignal carrying boundary, shown as rectangular, and further repeatedtraverses could produce yet more phases. In the example shown, thepositions A1, A2, B1 and B2 will yield localized four-phase clocksignals, as will the positions C1, C2, D1, and D2. The repeated boundarytraverses will be with suitable mutual spacing/separation of thetransmission-line 15 to avoid inter-coupling. FIG. 29 b shows idealizedfour-phase signal waveforms at points A1, A2, B1 and B2 and at C1, C2,D1 and D2.

FIG. 30 shows addition of an open-ended passive transmission-line (15 e,15 f) connected to the closed-loop transmission-line 15 and having thecharacteristics, of having an electrical length of 180.degree., ofproducing no adverse effect at the tap point, since it acts as anopen-circuit oscillating stub. Amplifiers 21 will not be present alongthis open-ended line 15 e,f but inverters 23 could be far ends of eachof the traces 15 c and 15 d to reduce risk of spurious oscillations.Indeed, tuned oscillation in such stubs 15 e,f can have usefulregenerative effects for the transmission-line 15 and thus serve forreinforcement and/or stability purposes.

Passive transmission-line connections with no particular requirement forimpedance matching can be used to connect oscillating transmission-linesof the same, or substantially the same, frequency together, at leastprovided that enough inter-connections are established between twosystems, at connection positions with the same relative phases in theinter-connected networks. Such connections can assist in synchronizinghigh speed digital signals between IC's and systems because non-clocksignals (i.e. the IC/system data lines) will have similar delaycharacteristics if they are incorporated into the same routing (e.g.ribbon cable, twisted pair, transmission-line) as the clock connections,thus making data and clocking coherent between different systems.

FIG. 31 shows one example of coherent frequency and phase operation oftwo clock distribution networks of two monolithic ICs 68.sub.1, 68.sub.2each having a clock generation and distribution hereof and pairs ofinter-IC connections E, F and G, H. The two ICs concerned will operatecoherently, i.e. at the same frequency and with the same phaserelationships, where each of the connections is substantially of180-degrees electrical lengths, or a multiple satisfying 360.degree . .. n+180.degree. where n is zero or an integer.

A single pair of inter-IC connections (E, F or G, H) will result infrequency and phase ‘locking’. More than one pair of inter-ICconnections (E, F and G, H as shown) will result further in clock wavedirection or rotation locking.

Also shown in FIG. 31 is a first and second ‘stub’ connections 82 and83, though there could be more of either or each. The first stubconnection 82 has a total electrical length of 180.degree. to assist instabilizing operation. The second stub connection 83 is open-ended andalso of 180.degree. electrical length and helpful for stabilization.Such stubs 82, 83 can be particularly useful for non-IC applications ofthe invention where conductive trace definition may be less precise thanfor ICs.

Impedance of the pairs of connections E, F and G, H and connections 82,83 can have any value since, in normal operation and once theseconnections are energized, there will be no net power flow therein forcorrect phasing thereof. It is, however, preferred that the impedance ofthese connections E, F and G, H and 82, 83 is greater than that ofoscillator transmission-lines 15 to which they are connected. Theseconnections will support a standing EM wave rather than a traveling EMwave.

Such FIG. 31 inter-connections can be applied equally well to intra-IC,inter-IC, IC-to-PCB and/or any non-IC, i.e. PCB-to-PCB systemconnections.

FIG. 32 illustrates digitally selectable shunt capacitors that areformed out of mosfet transistors.

Digitally selectable shunt capacitors illustrated in FIG. 32 can beoperatively connected to the transmission-line 15 and controlled for thetraveling EM wave to be delayed slightly, i.e. the frequency ofoscillation can be controlled. Such delays are useful for fine tuningthe frequency of a transmission-line(s). As shown, eight shuntcapacitors are implemented by means of mosfet transistors. The mosfetstransistors M1, M2, M5 and M6 are PMOS transistors and mosfettransistors M3, M4, M7 and M8 are NMOS transistors.

The mosfets M1, M3, M5 and M7 have their drain and source terminalsconnected to the ‘inner’ transmission-line conductor 15 a, for example,and the mosfets M2, M4, M6 and M8 have their drain and source terminalsconnected to the ‘outer’ transmission-line conductor 15 b. The substrateterminals of mosfets M1, M2, M5 and M6 are connected to the positivesupply rail V+ and the substrate terminals of mosfets M3, M4, M7 and M8are connected to the negative supply rail GND.

The gate terminals of mosfets M1 and M2 are connected together andcontrolled by a control signal CS0 and the gate terminals of mosfets M3and M4 are connected together and controlled by the inverse of controlsignal CS0. Likewise, the gate terminals of mosfets M5 and M6 areconnected together and controlled by a control signal CS1 and the gateterminals of mosfets M7 and M8 are connected together and controlled bythe inverse of control signal

The following truth table illustrates which mosfet shunt capacitors(M1-M8) contribute capacitance, i.e. ‘Mosfets On’, to thetransmission-line 15.

CS0 CS1 Mosfets ‘On’ Mosfets ‘Off’ 0 0 M1–M8 — 0 1 M1–M4 M5–M8 1 0 M5–M8M1–M4 1 1 — M1–M8

It is preferred that the respective sizes and numbers of shuntcapacitors connected to the ‘inner’ and ‘outer’ transmission-lineconductive traces 15 a, 15 b are the same, i.e. balanced. Whilst eightmosfet shunt capacitors M1–M8 are shown, any number of mosfet shuntcapacitors having suitable sizes, and hence capacitances, can be used,provided that the transmission-line 15 is balanced, as per FIG. 33.

There are other configurations for producing digitally controllableshunt capacitors that, may or may not be formed using mosfettransistors. One known example, again using mosfets, could be the use ofbinary weighted mosfet capacitors for example. Alternatives to MOScapacitors affording variable capacitance include varactors and PINdiodes for example.

It can be advantageous for the ‘capacitor arrays’ to be replicated atregular intervals around the transmission-line(s) so as to distributethe impedance.

The possibility is envisaged of achieving highest possible operatingfrequencies consistent with disconnectable switching of logic circuitry,including as semiconductor fabrication technology is bound to develop.

Indeed, transmission-line formations themselves should scale with ICprocess technology, thus smaller and faster transistor formations leadnaturally to shorter and faster transmission-line oscillators for yethigher clock frequencies.

Other possibilities include maintaining low power consumption;regardless of applications, which could be as to any resonating ofcapacitive and inductive connections to a transmission-line, andspecifically use relative to such as shift registers or‘precharge’/‘evaluate’ logic.

Whilst there is evident advantage in not having to use external timingreference such as a quartz crystal, nor PLL techniques, there may besituations and applications where this invention is applied inconjunction with such external timing crystals etc.

Turning to FIG. 34, signal paths 115, 215, 315 are shown of atransmission line nature, specifically also of the parallel dualconductive component/trace type shown previously for FIG. 1, see a,bsubscripting of 115, 215, 315.

Each of these transmission lines 115, 215, 315 has regenerative activemeans between its conductive traces, see bi-directional invertingswitching/amplifying circuitry 121 shown between the traces 315 a,b butonly for the line 315 and only once therefore to avoid cluttering thedrawing. As for FIGS. 1–33 embodiment, the circuitry 121 will be pluraland distributed preferably substantially evenly along each of thetransmission lines 115, 215, 315 in numbers and spacings affordingoperational effectiveness, up as many and as small as reasonablypractical.

The transmission line signal paths 115, 215, 315 carry arrow-headsindicating unidirectionality of signal flow therein, and thesesignal/flow directions are different as between next adjacent pathsspecifically opposite from right-to-left and left-to-right sequentiallyup and down the drawing. These directions of signal flows could comefrom opposite end (left/right) application of a drive signal (101), or(as shown) result from optional loop connection links 116 from one path(see 115) receiving the drive signal to the next path (see 115–215) andonwards as desired (see 215–315), say in groups of signal paths (see115–315) each with one path driven and others linked to achieve asuccessively contra-flow effect as illustrated. The last signal path sofed, whether of a group or overall, is terminated (see 117 for path315). These links 116 are shown as being of passive loop connectingnature, as is generally adequate to their purpose. It is to beappreciated that what is shown in FIG. 34 is typically fragmentary of amuch larger overall array, see dashed “etc” lines.

Importantly, a signal path with one direction of signal flow hascross-connection couplings to at least one other signal path withanother direction of signal flow, see 118 between paths 215 and 315having opposite unidirectional signal flows. As shown, these couplings118 are at localized adjacencies of the signal paths concerned and willbe of a non-linear active nature, say of switching transistor type orinverter type and advantageously bidirectional as specifically shownwith inverters in back-to-back configuration.

The illustrated active nature of the cross-connections 118 usefullystrengthens inter-coupling of the contra-flow signal paths concerned,including for gating signal flow energy to and from voltage sources andfor mutual energy exchange, in fact generally supplies energycontributory to maintaining desired operation. The spacings of thecross-couplings 118 is further contributory to desired operation,specifically being at substantially equal electrical length intervalsalong each signal path that have prescribed phase correlation to signalflows in such paths, see bracketed phase numbers at 90.degree.intervals; and positions that reinforce phase correlations between thepaths, see correspondence of bracketed phase numbers at thecross-connections 118.

The localized adjacencies, thus the cross-couplings 118, are shown at180 degree phase intervals along the paths 115, 215, 315. Thecross-couplings 118 are shown as bidirectional active nature,specifically back-to-back inverters much as for in-path connections 116,and in pairs (118 a,b) between the transmission line conductor traces(see 215 a,b) and 315 a,b) of the different signal paths concerned (215,315).

These bidirectional component conductor connections 118 a,b are betweenthe “a” conductors of one transmission line signal path and the “b”conductors of the other, respectively. This has the effect, for mutualenergy transfer between the paths, of affording “cross-over” effects,thus usefully effectively recirculatory Moebius-twist signal paths thatafford some degree of sustaining oscillation effect/action. TheMoebius-twist signal paths comprise one part from one signal path andanother from the other signal path, see for example between electricalphase positions (240, 60) and (150, 330) of the path 315, then (330,150) and (60, 240) of the path 215.

This phase-locking will not be as strong, nor as power efficient as forthe hard-wired connections of FIGS. 16 and 17, but there is in-principleviability for very fast and synchronized timing signal distributed oversubstantial areas without recourse to problematic H-tree distributionlay-out design; and as shown, inherently of differential natureaccording to input drive signal applied at 101, typically of square-waveform with its edges usefully maintained and refreshed by the crossconnections 121.

Also, as for FIGS. 1–33, there is oscillation without resonance, i.e.repeating periodicity related to signal path traversal time and arequisite degree of feedback via the interconnections 118 (though lessthan for the hard-wiring of FIGS. 16, 17); inherent rotation-locking ofsignal flows; absence of in-series one-way amplifier provisions andtheir requirements for specific inputs and outputs; and phase-lockingmore simply than available by such as servo control action inherent insuch as phase-locked loops. Moreover, there is significant energyconservation compared with such as H-tree distribution provisions asmany more loads can be served for each energy-absorbingreflection-limiting termination 117 (though again less than for theelectromagnetically continuously endless elements of FIGS. 1–33). Also,there will be useful energy exchanges with power supplies.

Turning to FIGS. 35 a and 35 b an endless electromagnetically continuousrecirculatory signal path 415 is shown in FIG 35 a, again of atransmission line nature, specifically comprising dual parallelcomponent conductors/traces (415 a,b), but now without a Moebius twist,thus without a cross-over or transformer to afford inversions. This isagain shown reliant on drive signal 401, thus effectively as a means todistribute such timing signal through or about a localized area; and, asfor all embodiments of this invention, is not reliant on any particulargeometry, whether or area services or the signal path itself.

FIG. 35 a has arrow heads showing a particular rotation direction forsignal flow round the endless path 415, as can be imposed and maintainedby illustrated active application of different phases of the drivesignal (401), specifically for a differential timing signal continuouslyrotating round the path 415 with its opposite phases in the componentpaths 415 a,b respectively, see inverters 402. Three phases of the drivesignal 401 are shown at 120-degrees intervals (60, 180, 300), and theconnection positions to the signal path 415 correspond in the context ofthat signal path 415 having an electrical length matching traveling wavefull rotation time with the full 360-degree period of the drive signal401.

Electrical energy for recirculatory signal flow round the path 415 isprovided by active amplifying action in the application of the drivesignal phases, see inverting coupling amplifiers 416, 417 and 418. Theseamplifiers 416, 417, 418 are in pairs subscripted a,b at each drivesignal connection position, one to each component trace 415 a,b inapplication and distribution of differential timing signals, so thatopposite or inverted timing signals are available in the two separatelyendless component conductors 415 a and 415 b at any take-off position,see at 420 with respective phasing (135, 315).

Relative to FIG. 35 a non-differential (or single-ended) operation is,of course, readily available by omission of the loop conductor/trace 415b and the drive signal connections through inverters 402 and amplifiers416 b, 417 b and 418 b.

Whilst detailing herein has been within the context of currentlydominant CMOS technology for ICs, it will be appreciated by thoseskilled in the art that principles are involved that are also applicableto other semiconductor technologies, e.g. Silicon-Germanium (Si—Ge),Gallium-Arsenide (Ga—As) etc.

Finally, highly beneficial particular utility in overcoming the problemsassociated with high frequency clocking, e.g. where F>1 GHz, no otherapplicability of combined timing signal generation and distribution isto be excluded from intended scope hereof, say for systems and apparatusto operate at frequencies less than 1 GHz.

1. A clock distribution system comprising: a plurality of transmissionline segments, each segment having ends and a length of spaced apartfirst and second conductors therebetween, each length of conductor beingelectrically continuous; a plurality of passive connection meanscoupling the ends of the one or more segments to form a serpentinesequence of segments, the serpentine sequence being configured toprovide positions at which adjacent segments are physically proximateand phase-correlated, the open end of the first segment in the sequencefor receiving an oscillator that provides a pair of oppositely phasedclock signals, the open end of the last segment in the sequence forconnecting to a load device; a plurality of regeneration devices locatedat various spaced-apart positions on each of the segments and connectedbetween the first and second conductors of the segment; and a pluralityof regeneration device pairs, each located at one of the physicallyproximate, phase-correlated positions between adjacent segments, whereina first one of the pair is connected between the first conductor of onesegment and the second conductor of the adjacent segment and a secondone of the pair is connected between the second conductor of the onesegment and the first conductor of the adjacent segment, and wherein,when driven by the oscillator, adjacent transmission line segments haveunidirectional waves traveling in opposite directions.
 2. A clockdistribution system as recited in claim 1, wherein each regenerationdevice includes a first inverter and a second inverter, each having aninput and an output, the input of the first inverter connected to theoutput of the second inverter and the input of the second inverterconnected to the output of the first inverter.
 3. A clock distributionsystem as recited in claim 1, wherein each of the regeneration devicepairs, wherein each regeneration device pair includes two regenerationdevices; and wherein each regeneration device includes a first inverterand a second inverter, each having an input and an output, the input ofthe first inverter connected to the output of the second inverter andthe input of the second inverter connected to the output of the firstinverter.
 4. A clock distribution system as recited in claim 1, whereineach of the passive connection means includes a pair of wire links.
 5. Aclock distribution system as recited in claim 1, wherein the pluralityof regeneration devices is nearly evenly distributed along each of thesegments.
 6. A clock distribution system as recited in claim 1, whereinthe load device is resistive.
 7. A clock distribution system as recitedin claim 1, wherein each of the segments has an impedance; and whereinthe load device has an impedance that is matched to the impedance of thelast segment.
 8. A clock distribution system as recited in claim 1,wherein phase correlated positions between adjacent segments are 60degrees on one segment and 240 degrees on the adjacent segment.
 9. Aclock distribution system as recited in claim 1, wherein phasecorrelated positions between adjacent segments are 90 degrees on onesegment and 270 degrees on the adjacent segment.
 10. A clockdistribution system as recited in claim 1, wherein phase correlatedpositions between adjacent segments are 180 degrees on one segment and 0degrees on the adjacent segment.
 11. A clock distribution system asrecited in claim 1, wherein phase correlated positions between adjacentsegments are 330 degrees on one segment and 150 degrees on the adjacentsegment.